INTERNAL DUAL-BAND PATCH ANTENNA FOR MOBILE PHONES Jani Ollikainen, Outi Kivekäs, Anssi Toropainen, and Pertti Vainikainen Helsinki Universit of Technolog, Institute of Radio Communications, Radio Laborator, P.O. Bo 3, FIN-21 HUT, Finland Email: jol@radio.hut.fi INTRODUCTION Development of small antennas for mobile phones has received a lot of attention during the last few ears due to sie reduction of the handsets, requirements to keep the amount of RF power absorbed b the user below standardied levels regardless of the handset sie, and introduction of multimode phones. Single-band and dual-band phones in which the eternal whip or heli antenna has been replaced b an internal shorted patch antenna (or PIFA) are alread in the market. It ma be assumed that in the future there will also be a need for internal antennas which operate in several communication sstems. In this paper, a multiresonant dual-band internal handset antenna is introduced. The antenna is based on the dual-resonant antenna structure reported in [1]. To our knowledge, this is the first internal handset antenna which covers the E-GSM9 (88 MH - 96 MH), GSM18 (171 MH - 188 MH), DECT (188 MH - 19 MH), PCS19 (18 MH - 199 MH), and UMTS (19 MH - 217 MH) frequencies with a return loss L retn 6 db and high radiation efficienc. Previousl shorted patch antennas (or PIFAs) with more than one band of operation have been reported e.g. in [2-4]. The dual-band PIFA was introduced in [2]. The idea was etended to multiband PIFAs in [3]. A compact dual-band PIFA with two capacitive feeds was reported in [4]. ANTENNA DESIGN The studied antenna consists of three shorted patches, one for the lower band and two for the upper band. The lower band patch (parts a-c in Fig. 1) and one of the upper band patches (part d) have been joined together to form a dual-band element having onl one short circuit and one feed. This dual-band element has been folded into a meander line -like shape to better fit it into the geometr of a mobile phone. The third shorted patch (part e) is positioned net to the upper band section of the dual-band element to obtain a doubl tuned resonance and a wider impedance bandwidth at the upper band. The dielectric between the patches and the ground plane is air (relative permittivit ε r = 1). The antenna is fed b a probe which is connected to the upper band patch of the dual-band element. The total sie of the antenna element is 4 mm 3.4 mm 7.2 mm (width length thickness). It is positioned on top of a ground plane having dimensions 4 mm 11 mm.3 mm (width length thickness). The length and width of the ground plane were selected to be approimatel equal to those of the printed circuit board of a tpical mobile phone. The antenna structure was designed and theoreticall studied using IE3D which is a moment method-based 3D electromagnetic simulation program b Zeland Software. A prototpe antenna was constructed b photoetching the patches, short circuits and the capacitive load from.2 mmthick sheet of tin brone (contains 96 % copper and 4 % tin, conductivit σ = 1.1 1 7 S/m). The ground plane was made out of.3 mm-thick sheet of the same material. The patches are supported above the ground plane b the short circuits, the feed probe, and a 4 mm 1 mm 7 mm piece of Strofoam. Acknowledgements - The postgraduate studies of the first two authors have been financiall supported b the Academ of Finland, Graduate School in Electronics, Telecommunications and Automation (GETA), Jenn and Antti Wihuri Foundation, Nokia Foundation, and Tekniikan Edistämissäätiö. The radiation patterns of the constructed prototpe antenna were measured in the anechoic chamber of LK-Products, Finland.
2 2 4.3 7 4 Short circuit Feed a b c d e 3. 1. 1. 4 11 Capacitive load 1. 1. Part Length (mm) Width (mm) a 3 9 b 3 6 c 3 6. d 29 6. e 28.8 6 Fig. 1. Studied antenna configuration. All dimensions are in millimeters. RESULTS The frequenc responses of the measured and simulated reflection coefficients are shown in Fig. 2. A good agreement between the simulated and measured results can be observed. The measured impedance bandwidth (L retn 6 db) of the lowest band is 12.4 % (11 MH) at the center frequenc f c = 926 MH. For the dual-resonant upper band, the measured bandwidth (L retn 6 db) is 27.7 % (3 MH) at f c = 1934 MH. Above the desired bands, there is also a third band with the bandwidth of 3.6 % (89 MH) at f c = 21 MH. The third band is assumed to be caused b the 3λ/4-resonance of the lower band section of the meandered antenna element. This resonance has been tuned down from 27 MH b the presence of the upper band section of the meandered dual-band element (part d in Fig.1). According to simulations, the impedance bandwidth of the antenna depends on the length of the ground plane. However, in this design the ground plane length is not optimied with respect to the bandwidth. S11 (db) 3 6 9 12 1.7 GH 2.7 GH 1.4 GH 1.4 GH 18 21 24 Measured Simulated 27.7.9 1.1 1.3 1. 1.7 1.9 2.1 2.3 2. 2.7 Frequenc (GH) Fig. 2. Measured (solid line) and simulated (dashed line) reflection coefficient as a function of frequenc. Dotted circles on the Smith charts represent L retn = 6 db. The measured and simulated cuts of the radiation patterns of the studied antenna are shown in Fig. 3. The presented cuts have been obtained in free space. At 92 MH, the radiation pattern is almost omnidirectional and resembles that of a half-wave dipole. At 171 MH, the pattern becomes more comple and somewhat more directive. The directivit increases with frequenc. At the same time the pattern maimum tilts towards the negative -ais. In addition, it can be seen in Fig. 3 that the polariation of the antenna at the lower band is different from the polariation at the upper band. Fig. 4. shows the simulated radiation efficienc (IE3D) of the antenna in free space. The conductivit used in the calculations was σ = 1.1 1 7 S/m. The efficienc stas above 9 % over the two desired bands of operation.
f = 92 MH 9 3 6 1 1 2 2 3 3 6 9 9 6 3 1 1 2 2 3 3 6 9 9 6 3 1 1 2 2 3 3 6 9 12 12 12 12 12 12 1 18 1 1 18 1 1 18 1 f = 171 MH 9 3 6 1 1 2 2 3 3 6 9 9 6 3 1 1 2 2 3 3 6 9 9 6 3 1 1 2 2 3 3 6 9 12 12 12 12 12 12 1 18 1 1 18 1 1 18 1 f = 217 MH 9 3 6 1 1 2 2 3 3 6 9 9 6 3 1 1 2 2 3 3 6 9 9 6 3 1 1 2 2 3 3 6 9 12 12 12 12 12 12 1 18 1 1 18 1 1 18 1 Fig. 3. Measured (o o o E θ, E ϕ ) and simulated ( E θ, E ϕ ) cuts of the radiation patterns at 92 MH, 171 MH, and 217 MH. Antenna orientation is given in Fig. 1. Results include the effect of mismatch loss. Radial unit is dbi. 1 Radiation efficienc (%) 9 9 8 8 7 7.7.9 1.1 1.3 1. 1.7 1.9 2.1 2.3 2. 2.7 Frequenc (GH) Fig. 4. Simulated (IE3D) radiation efficienc in free space as a function of frequenc for the studied antenna. Conductivit of metal parts σ = 1.1 1 7 S/m.
The specific absorption rate (SAR) characteristics of the studied antenna were evaluated b simulations which were performed using a commercial FDTD program (XFDTD, version. Bio-Pro b Remcom, Inc.). An FDTD mesh of a male head and shoulders with the voel resolution of 2. mm was used. This was based on the standard human head and shoulders mesh (3 mm voel) obtained from Remcom. The values of the tissue parameters (ε r, σ eff ) were interpolated from the values given in []. The tissue densit values were those provided with the original mesh. The phone model (antenna and thin ground plane) was placed beside the head model according to the intended use position specified b CENELEC [6]. This was obtained b rotating the head model forward b 74 and to the right b 1 while keeping the phone model vertical in the mesh. The distance of 11.9 mm ( cells) was left between the head and phone model. The SARs were calculated at 91 MH, 173 MH, and 21 MH using a stead-state sinusoidal ecitation. Due to the limitations of the 2. mm grid, the dimensions of the antenna, and thus also its resonant frequencies, were slightl different from those of the measured antenna. The frequencies used in SAR simulations were adjusted accordingl. The simulation space enclosing the head and the handset consisted of 27 187 26 cells. The simulations were run for 4 timesteps to ensure converged results. The results of the SAR simulations are listed in Table 1. The maimum 1 g average SAR value was located near the center of the ground plane at 91 MH. At 173 MH, the maimum was found near the short circuit of the parasitic resonator (part e in Fig. 1), and at 21 MH near the short circuit of the meandered patch. The results indicate that the SARs are below the basic restrictions set b CENELEC. When studing the SAR values given in Table 1, it must be noted that the specified mean (rms) output powers for handsets in GSM9, GSM18, DECT, and UMTS sstems are 2 mw, 12 mw, 1 mw, and 2 mw, respectivel. The radiation efficiencies when the phone model is beside the head are also listed in Table 1. The radiation efficiencies at 173 MH and 21 MH are more than twice as high as the efficienc at 91 MH. This is partl eplained b more directive radiation patterns at 173 MH and 21 MH. Table 1. Simulated maimum 1 g average SAR values in the head, and simulated radiation efficiencies (XFDTD) with prototpe beside the head. SAR results are normalied to 1 W of CW input power. f (MH) 91 173 21 Ma 1 g average SAR (W/kg) 4.6 1.8 1.8 Radiation efficienc (%) 29 72 7 CONCLUSIONS A multiresonant internal dual-band shorted patch antenna for mobile phones has been studied. Both simulated and measured results have been presented. The studied antenna has two bands of operation, the lower one ranges from 868 MH to 983 MH (12.4 %) and the upper one from 1666 MH to 221 MH (27.7 %). The radiation patterns are suitable for internal handset antenna application. According to simulated results with FDTD, the SAR values caused b the antenna are below the limits set b CENELEC. REFERENCES [1] J. Ollikainen and P. Vainikainen, Radiation and bandwidth characteristics of two planar multistrip antennas for mobile communication sstems, Proc. IEEE 48th Vehicular Technolog Conference, Vol. II, Ottawa, Ontario, Canada, Ma 18-21, 1998, pp. 1186-119. [2] Z. D. Liu, P. S. Hall, and D. Wake, Dual-frequenc planar inverted-f antenna, IEEE Transactions on Antennas and Propagation, Vol. 4, No. 1, October 1997, pp. 141-148. [3] P. Song, P.S. Hall, H. Ghafouri-Shira, and D. Wake, Triple-band planar inverted-f antenna, IEEE Antennas and Propagation International Smposium Digest, Vol. 2, Orlando, Florida, Jul 11-16, 1999, pp. 98-911. [4] C. R. Rowell and R. D. Murch, A compact PIFA suitable for dual-frequenc 9/18-MH operation, IEEE Transactions on Antennas and Propagation, Vol. 46. No. 4, April 1998, pp. 96-98. [] User s Manual for XFDTD the Finite Difference Time Domain Graphical User Interface for Electromagnetic Calculations, Version.4, Remcom, Inc., Februar 1999, 128 p. [6] European specification (ES 9), Considerations for the Evaluation of Human Eposure to Electromagnetic Fields (EMFs) from Mobile Telecommunication Equipment (MTE) in the Frequenc Range 3 MH - 6 GH, Brussels, Belgium, CENELEC, October 1998, 81 p.
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