EuMC: A Ka-Band RF-MEMS Phase Shifter Approach Based on a Novel Dual-State Microstrip Line

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1 Proceedings of the 37th European Microwave Conference A Ka-band RF-MEMS phase shifter approach based on a novel dual-state microstrip line C. Siegel #1, V. Zieglerl #2, U. Prechtel #3, B. Schönlinner #4, H. Schumacher *5 # EADS Deutschland GmbH, EADS Innovation Works Germany, SI-MW, Munich, Germany 1 christian.siegel@eads.net 2 volker.ziegler@eads.net 3 ulrich.prechtel@eads.net 4 bernhard.schoenlinner@eads.net * University of Ulm, Dept. of Electron Devices and Circuits, Albert Einstein Allee 45, 8981 Ulm, Germany 5 hermann.schumacher@e-technik.uni-ulm.de Abstract This paper presents the design and realization of a 3-bit RF-MEMS based phase shifter at 34GHz using a very low complexity and highly reliable RF-MEMS technology on silicon. The three bits of the circuit use different techniques to achieve the necessary phase shifts. One new design technique, the dual-state microstrip line, is enabled by the presented RF-MEMS technology and changes the effective ε r of a microstrip transmission line by lifting part of it into the air. This leads to a change in the electrical length of the microstrip transmission line, which in turn results in a phase shift. The related insertion loss of the 45 bit is less then -.35dB and a matching better -19dB for both switching states. Further on, a loaded-line bit with 45 of phase shift is realized by switching between a capacitive and an inductive load. The capacitive switching state shows an insertion loss of -.4dB and a matching better -13dB, while the inductive load has an insertion loss of -.7dB and a matching of -25dB. The loaded-line bit combined with the dual-state microstrip line is used for the 9 bit. An additional miniaturized switched line phase shifter is implemented for the 18 state. The three bits were combined and measured in all states. The results of the 3-bit phase shifter are shown with a mean insertion loss of -2.2dB and a phase derivation of at 34GHz. I. INTRODUCTION In modern radar and communication systems, there is a growing need for high performance, but low cost microwave phase shifters. They are used to build phased array antennas, which are besides the electronically beam steering capable of suppressing certain spatial directions [1], [2]. This mode can be used to fade out other disturbing communication or radar systems. RF-MEMS phase shifters are excellent candidates to be used in phased array antennas. They have shown very good performance in microwave applications [3]-[4] in terms of low insertion loss and high linearity. In addition, their very low power consumption allows a high element density without special thermal considerations. But today, still some challenges for RF-MEMS exist and delay their commercial use. The main issues being their reliability under RF-power and the rather complex packaging compared to semiconductor devices. In this paper, a new phase shifter approach is presented that contains a dual-state microstrip transmission line and RF- MEMS switches to build standard phase shifters. II. RF-MEMS TECHNOLOGY AND ITS BENEFITS The RF-MEMS devices presented in this paper consist of a singly clamped metal cantilever that rolls up due to an intrinsic stress gradient in the metal. The cantilever can be switched onto the silicon substrate with electrostatic actuation. The metal cantilever consists of only one aluminum alloy to allow a high temperature range of operation. The switches and the fabrication process were described in an earlier work [5]. They are realized using a highly doped silicon region instead of a second metallization layer to keep the high quality thermally grown silicon oxide as dielectric layer. Due to the rolling design, the absolute height of the tip of the cantilever is quadratic proportional to the length of the cantilever. A typical height of a 3µm long beam is in between 2µm and 25µm. The actuation voltage is mainly depending on the stress gradient and the thickness of the metal beam and the sacrificial layer. The last one determines the distance between substrate and cantilever at its anchor point. The benefit of this technology beside the excellent RFperformance is clearly the low complexity of process to realize the dual-state transmission lines or the switches. Thus the devices can be fabricated at low costs and have additionally demonstrated a very high reliability. The switches realized in this technology achieved lifetime cycling in excess of 19 switching cycles (duty-cycle 5%, switching frequency 5Hz) and an operation temperature up to 12 C [6]. III. DUAL-STATE MICROSTRIP LINE PHASE SHIFTER The principle of this novel approach is to achieve a phase shift by lifting parts of a microstrip line off the substrate. Thereby, the effective ε r and with it the electric length of a microwave line gets smaller. The signal on the dual-state transmission line gets a larger phase shift in the down-state and a smaller one in the up-state. In general, the amount of the phase shifting effect depends on the average change of r eff of the whole line EuMA 1221 October 27, Munich Germany

Previously, a microstrip line phase shifter realized by switching up on each side 1µm of the width of a 26µm wide microstrip transmission line was shown [7].

In the actual design, where all the line is switchable and is only fixed to the substrate with small additional stubs, a phase shift of 15 /mm at 34GHz is reached.

Therefore, the overall loss on the transmission line is also a function of the applied DC-voltage that is necessary for switching.

2 Previously, a microstrip line phase shifter realized by switching up on each side 1µm of the width of a 26µm wide microstrip transmission line was shown [7]. This resulted in a phase shift of 5 /mm of line length. By fixing the line on one side and thus switching 2µm on the other side, increases the tuning effect to 12 /mm. In the actual design, where all the line is switchable and is only fixed to the substrate with small additional stubs, a phase shift of 15 /mm at 34GHz is reached. The losses of a microstrip line on silicon are mainly dominated by the remaining conductivity of the substrate and by a conductive channel on the interface of the oxide and the silicon [8]. Therefore, the overall loss on the transmission line is also a function of the applied DC-voltage that is necessary for switching. Since in the up-state, no voltage is applied, the line loss is lower than in the down-state. The line loss without applied voltage on the high resistivity silicon with thermal oxide is about.1db/mm at 34GHz. The insertion loss at 34GHz is below -.36dB in both states and the achieved phase shift is 45 (Fig. 3). Phase in up 9 down phase shift Fig. 3 Phase shift of the dual-state microstrip line phase shifter. IV. LOADED LINE BIT Another often used approach to realize a phase shift is to variably load a transmission line. The load is in this case switched between capacitive (switches up) and inductive (switches down). The inductive load is a stub that is a little bit shorter than λ/2. For matching reasons, there are two loads with an inductive line region in between at a distance of λ/4. The switches are actuated by the thin aluminum lines, which are decoupled at the design frequency by the radial stubs at a distance of λ/4 (Fig. 4). Fig. 1 SEM picture of the dual-state microstrip line From the circuit design point of view, the impedance of the line increases with decreasing of ε r eff in the up-state. For a minimal insertion loss of the phase shifter however both states have to be matched. The down state is designed to be a 5Ohm line whereas the up-state is matched by choosing its electrical length equal to λ. At the frequency, where the length of the movable part is λ/2, the line is also matched. For the dual-state microstrip line both states are matched better -19dB (Fig. 2). S21 in db Fig. 2 Insertion loss and matching of the dual-state microstrip line phase shifter black: up-state, green: down-state S11 in db Fig. 4 left: Photography of the loaded line phase shifter, right: Photography of the miniaturized 18 switched line phase shifter. The measured S-parameters are shown for both loading states in the following Figure 5. S21 / S11 in db 5 S21 Capacitive -5 S21 Inductive S11 Capacitive -2 S11 Inductive Fig. 5 Insertion loss and matching of the loaded line phase shifter The insertion loss is.4db for the capacitive load with a matching better 13dB and.7db with a matching better 1222

2dB for the inductive load at 34GHz. The related phases of both switching states are depicted in Figure 6. The phase shift achieved at 34GHz is 43. Phase in -22.5-45 -67.5 Inductive, down -9-112.

3 2dB for the inductive load at 34GHz. The related phases of both switching states are depicted in Figure 6. The phase shift achieved at 34GHz is 43. Phase in Inductive, down Capacitive, up Fig. 6 Measured phase for both switching states. Squares indicate the up-state. V. MINIATURIZED SWITCHED-LINE 18 -BIT The loaded line approach is most suitable for phase shifts up to 9, but tend to consume a lot of space and to have higher insertion losses for larger phase shifts. For these reasons, the miniaturized 18 bit was designed in a switched line configuration (Fig. 4 right). It has a size of 16µm in length and 18µm in width (including two times the substrate thickness next to the microstrip lines). For the phase shifting, one can either close the upper switch to reach of phase delay or the lower switch to reach a phase delay of 18. If the upper switch is in down-state, the lower switch is open and acts as a loading stub as the other open-ended line does. Both loads are in this case capacitive ones and their distance is chosen that the reflections get partly compensated. S21 in db S11 in db Fig. 8 Simulated and measured matching of the switched line phase shifter for both states. VI. 3-BIT PHASE SHIFTER In order to realize a 3-bit phase shifter, two dual-state line bits, one loaded line bit and the switched line bit are combined (Fig. 9). The different bits are separated by interdigital decoupling capacitors. The DC-actuation is applied to the switches via high resistivity carbon lines. All title and author details must be in single-column format and must be centered. Fig. 9 Picture of the 3-bit phase shifter with carbon lines for biasing of the different switching states. The 9 bit is a combination of a 45 dual-state transmission line and a 45 loaded line bit. The 45 bit is another dual-state transmission line. The overall size of the 3- bit phase shifter is 1mm * 4mm. The Figures 1 to 12 show the measured insertion loss, the return loss and the phase for all eight switching states Fig. 7 Simulated and measured insertion loss of the switched line phase shifter for both states. The insertion loss for both states of the switched line phase shifter is shown in Figure 7. The measured loss is 1.2dB for the state with a matching of 11dB (Fig. 8). This fits well with the simulations. The 18 state shows losses of 2.2dB with a matching of 1dB. The simulation shows much lower loss. A possible explanation is radiation induced losses from the two parallel line sections. The current in these lines has a phase difference of 18, thus the corresponding magnetic field will enclose both of them. Fig. 2 Measured insertion loss of the 8 states of the 3-bit phase shifter The measured insertion loss at 34GHz for all states is between 1.2dB and 3.1dB. The return loss is between - 1dB and -15dB except for the and the 18 state. 1223

4 The phase behavior for the eight switching states is indicated in Figure 12. The phase shift at 34GHz is between 36.4 and 56. The phase behavior for the eight switching states is indicated in Figure 12. The phase shift at 34GHz is between 36.4 and 56. Mean Insertion Loss in db Phase Derivation in deg Fig. 13 Mean insertion loss (left y-axis) and phase derivation (right y-axis) of the 3-bit phase shifter derived from the measurements for the use in a phased array antenna. Fig. 31 Measured return loss for the 8 switching states On the left Y-axis of Fig. 13 the mean insertion loss of the eight switching states is derived from the measurements. It is found to be below 2.55dB from 32GHz to 36GHz with 2.2dB at the design frequency. VII. CONCLUSIONS The successful design, fabrication and characterization of a Ka-Band RF-MEMS phase shifter using a novel dual-state transmission line concept were presented. Using the very low complexity RF-MEMS process, a 3-bit phase shifter consisting of a loaded-line bit, a dual-state line bit and a miniaturized switched-line bit was introduced. The phase shifter exhibits a low mean insertion loss of -2.2dB and a very good phase derivation of at 34GHz. Fig. 42 Measured phase of the 8 switching states In a phased antenna array, many of these phase shifters are used in parallel in different phase states. The exact phase state of each antenna element that is needed to focus the antenna beam in a certain spatial direction can not be ideally set due to the digital character of the presented phase shifters. This phase error has to be minimal, since it causes a reduction of the overall antenna gain called quantisation loss, a degradation of sidelobe levels and a deviation in the pointing accuracy. The phase derivation, shown in Fig. 13 (right Y-axis), is commonly used as a measure for the phase errors. The phase derivation for an ideal 3-bit phase shifter is 13. The presented phase shifter has a phase derivation of less than 14.2 over the frequency range from 32GHz to 36GHz with a minimum of at 34GHz derivation calculates to an equivalent number of bits of 2.87, a pointing accuracy of 6.15, a sidelobe degradation of 17.3dB and a quantisation loss of.265db. For an ideal 3-bit phase shifter the pointing accuracy is 5.625, the sidelobe degradation is -18.6dB and the quantisation loss is.26db. REFERENCES [1] R. Vincenti Gatti, L. Marcaccioli, R. Sorrentino, "A novel phase-only method for shaped beam synthesis and adaptive nulling," 33rd EuMC, European Microwave Conference, Munich, Germany, Oct. 23. [2] L. Marcaccioli, R. Vincenti Gatti, R. Sorrentino, "Real-time interference nulling method for mobile communication systems," IEEE RAWCON 24, Radio and Wireless Conference, Atlanta, Georgia, USA, Sept. 24. [3] G. M. Rebeiz, J. B. Muldavin, RF MEMS switches and switch circuits, IEEE Microwave Mag., pp 59-71, December 21. [4] A. Abbaspour-Tamijani, L. Dussopt, G. M. Rebeiz, Miniatur and Tunable Filters Using MEMS Capacitors, IEEE Transaction on MicrowaweTheorie and Techniques, Vol. 51, No.7, pp , July 23. [5] C. M. Siegel, V. Ziegler, B. Schönlinner, U. Prechtel, H. Schumacher, "Simplified RF-MEMS Switches Using Implanted Conductors and Thermal Oxide", European Microwave Conference, Manchester UK Sept. 26. [6] A. Stehle, C. Siegel, V. Ziegler, B. Schönlinner, U. Prechtel, S. Thilmont, U. Schmid, H. Seidel, RF-MEMS Switch Optimized for High Temperature of Operation, submitted to EuMW 27, Munich, Germany, Okt. 27. [7] C. Siegel, V. Ziegler, U. Prechtel, H. Schumacher, Low-complexity RF-MEMS technology for microwave phase shifting applications, German Microwave Conf., pp.13-16, Ulm, Germany, April 25. [8] M. Spirito, et al, Surface Passivated High-Resistivity Silicon as True Microwave Substrate, IEEE Transaction on MicrowaweTheorie and Techniques, Vol. 53, No. 7, pp , July 25.S. M. Metev and V. P. Veiko, Laser Assisted Microtechnology, 2nd ed., R. M. Osgood, Jr., Ed. Berlin, Germany: Springer-Verlag,

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